Copyright © 1980 American Telephone and Telegraph Company
The Bell System Technical Journal
Vol. 59, No. 8, October 1980
Printed in U.S.A.
High-Speed Digital Lightwave Communication
Using LEDs and PIN Photodiodes at 1 .3 /im
By D. GLOGE, A. ALBANESE, C. A. BURRUS, E. L. CHINNOCK,
J. A. COPELAND, A. G. DENTAI, T. P. LEE, TINGYE LI,
and K. OGAWA
(Manuscript received April 9, 1 980)
At the wavelength of 1.3 /im, fiber loss and dispersion are suffi-
ciently small that many lightwave communications applications can
use simple and reliable led transmitters and pin photodiode receiv-
ers without avalanche gain. Bit rates can be as high as several
hundred Mb/s. With new high-speed devices based on III-V semi-
conductors and microwave silicon ic technology, we have designed
two fully retimed optical regenerators that operate at 1.3 [im and at
bit rates of 44.7 and 274 Mb/s to study the potential of the led -pin
approach. A detailed analysis of the baseband characteristics of the
led, the fiber, and the receiver leads to an overall equalization
approach that minimizes receiver noise. The success of this perform-
ance optimization is corroborated by bit- error measurements under
simulated system conditions. The results suggest repeaterless opera-
tion over distances up to 24 km at 44. 7 Mb/s and 8 km at 274 Mb/s
for a cable loss of 1 dB/km and a bandwidth of 1000 MHz -km. The
use of lasers in such multimode fiber sytems would permit larger
margin allocations and penalties than those chosen for led systems,
but would not lead to substantially longer repeater spacings.
I. INTRODUCTION
Optical-fiber communication systems that use leds as sources and
pin photodiodes as detectors offer the attractive features of reliability,
simplicity, and low cost. Operation of such systems at high digital
speeds is now made possible by the availability of InGaAsP leds and
detectors, and of low-loss fibers that exhibit minimal material disper-
sion in the wavelength region near 1.3 /xm. 1 The bandwidth-distance
product at the wavelength of minimum dispersion is inversely propor-
tional to the square of the spectral width of the source and, for
germania-doped silica fibers, is 3.5 GHz • km or more if surface-emitting
1365
leds are used. 2 Thus high-speed digital systems of a few hundred
Mb/s can be operated over repeater spans of 5 to 10 km.
An earlier paper 3 presented a discussion of material-dispersion-lim-
ited operation of fiber systems that use leds and reported an experi-
ment that showed the effects of material dispersion on a 137-Mb/s
data link operating at X = 0.89 or 1.20 jum. In the present paper, we
discuss further the application of leds and pin photodiodes in high-
speed digital fiber systems and describe two optical repeater experi-
ments (with bit rates of 44.7 and 274 Mb/s) at X - 1.3 /rai using
InGaAsP leds as transmitters and InGaAs pin photodiodes followed
by GaAsFET preamplifiers as receivers. Experimental results are com-
pared with theory.
II. SYSTEM CONSIDERATIONS
For high-speed digital applications, the characteristics of the led
that are important are its reliability, output power, modulation band-
width, spectral width, and temperature behavior. The reliability of
high-radiance, surface-emitting InGaAsP leds is now well established,
as results from accelerated-aging tests at elevated temperatures project
a mean life in excess of 5 X 10 9 hours for room-temperature operation. 4
Sufficient power at A = 1.3 jum is available from these devices: peak
powers of 200 /xW have been coupled into large-core step-index optical
fibers. 4,5 The modulation bandwidth of an led depends on the device
geometry, the current density, and the doping concentration and width
of the active region; a high doping concentration or a narrow active
region results in large bandwidth but yields low output power. 6
The fall-off of the modulated output power P s (f) of an led as a
function of the modulating frequency /can be described by the function
H a {f) m (1 + if/b)-\ (1)
where b is the modulation bandwidth of the led defined at the half-
power point (3-dB electrical). For high-radiance surface-emitting
diodes, typical values of the average power P s that can be coupled into
standard multimode fibers (50-jum core diameter, 0.23 n.a.) are P a = 50
juW for b = 50 MHz. In general, P a is proportional to b~" where cSl.
For moderately fast, double-heterostructure GaAlAs surface emitters,
v = % is a typical value. 6 leds with bandwidths in excess of 1 GHz
have been fabricated; 7 the output of these very fast diodes tends to
decrease as 1/6.
Compared to an injection laser, the output power of the led is
rather insensitive to temperature variations. Observed data on the
performance of the led used in our repeater experiments are given in
Section III.
Microwave GaAsFETs are ideally suited for use as low-noise pream-
1366 THE BELL SYSTEM TECHNICAL JOURNAL, OCTOBER 1980
plifiers following the pin photodetector in an optical receiver because
they have low input gate capacitances and high-transconductances. 8 ' 9
The sensitivity of an optical receiver using state-of-the-art pin photo-
diodes and GaAsFETs can be made to fall within a few decibels of that
of the typical silicon avalanche photodiode receiver used presently in
fiber systems that operate at X = 0.85 /xm. 10,11 At longer wavelengths,
significant bulk leakage currents at voltages approaching breakdown
severely limit the use of avalanche gain as a low-noise amplification
process. For this reason, very low-noise preamplifiers will play a key
role in future long-wavelength systems.
We make use of Personick's theory 12 to calculate the receiver sen-
sitivity, which can be expressed in terms of N r , the average number of
signal photoelectrons per bit required at the detector to yield an error
probability of 10~ 9 . N r is given by three terms which account for
Johnson, shot and fet channel noise:
qB [ R L g m _ J
1/2
(2)
where
q = electronic charge
B = bit rate
k = Boltzmann's constant
R L = equivalent input resistance
II = total leakage current (photodiode and fet)
r = noise factor associated with channel noise in the fet and is
taken to be 1.15 for a GaAsFET with negligible gate noise 1213
T = absolute temperature in degrees Kelvin
Ct = total capacitance (photodiode, fet and stray)
g m = transconductance of the fet.
The coefficients <Ji and J% account for the noise distribution in the
receiver channel and are discussed subsequently.
Figure 1 presents receiver sensitivity estimates using eq. (2), assum-
ing Rl to be infinite and the best available devices to have g m = 30 mS
and Ct = 1.5 pF. For II = 0, N r is proportional to Ct/ "fgin- The dashed
diagonal lines help to determine the sensitivity in terms of the average
optical power
P r = NrhcB/X-q (3)
required at the receiver, where h is Planck's constant and c is the
speed of light. They are computed for a wavelength A = 1.3 /xm and an
external quantum efficiency tj = 0.7.
Figure 1 also indicates the performance degradation resulting from
leakage or dark current in the fet and the photodiode. Such currents
HIGH-SPEED DIGITAL LIGHTWAVE COMMUNICATION 1367
g m = 30 mS
7J = 0.7
X = 1.3 uri
10 100
BIT RATE IN MEGABITS PER SECOND
1000
Fig. 1 — Average number of photoelectrons per bit vs bit rate according to eq. (2)
required for error probability of 10~ 9 ; optical receiver uses piN-GaAsFET front-end with
total input capacitance Ct, leakage current h, and transconductance g m = 30 mS.
Dashed lines indicate average optical power required at the receiver for quantum
efficiency tj = 0.7 and wavelength X = 1.3 /xm.
can be as high as 200 nA at typical elevated operating temperatures
(see Sections III and IV). Another noise phenomenon, 1//" noise, is not
indicated in Fig. 1, but is discussed briefly at the end of Section IV. A
similar performance evaluation on the basis of Personick's theory has
been conducted for receivers using the most advanced silicon bipolar
transistors (Ct = 2pF, current gain /? = 300). The results are listed in
Table I for some bit rates of interest. Also listed is the expected
performance for the case of an avalanche photodiode followed by a
GaAs fet amplifier. In this case, we assume a maximum bulk leakage
1 368 THE BELL SYSTEM TECHNICAL JOURNAL, OCTOBER 1 980
Table I — Comparison of expected sensitivity of
three types of receivers
Bit Rate 44.7 Mb/s 274 Mb/s 1096 Mb/s
PIN/(Si)Bipolar -45 dBm -38 dBm -32 dBm
PIN/(GaAs)FET -50 dBm -40 dBm -32 dBm
APD/(GaAs)FET -43 dBm -38 dBm
current of 200 nA before multiplication and a carrier-ionization ratio
of 4:1.
The use of J 2 and J 3 in eq. (2) is based on the assumption that the
digital information arrives undistorted at the receiver input and is
transformed by the receiver filter function H r (f) for minimum noise
and intersymbol interference. In this case,
Jn+2 'B
H r
df. (4)
Table II lists J n for a non-return-to-zero signal format and a raised-
cosine signal spectrum at the receiver output, in which case H r = {irf/
2B) cot (irf/2B).
In general, the signal arriving at the receiver has been modified by
the response of the source, H s (f), and the response of the transmission
medium, H t {f). To compensate for these effects, the receiver must be
designed with a response H' r = H r /H s H t . As a first example, consider
the case of an led with response given by eq. (1) and unlimited fiber
bandwidth. Let fet channel noise be the dominant noise contribution
as described by the third term in eq. (2). The coefficient J$ in that
term now becomes Ja + J a (B/b) 2 because H r in eq. (4) is replaced by
H'r. Hence the required receiver power becomes P' r = P r [l + (Ja/Jz)
(B/b) 2 ] l/2 where P r is the value previously computed from eqs. (2) and
(3). This degradation of receiver sensitivity represents the power
penalty one pays for equalizing the response of the led.
The available optical signal margin between transmitter and receiver
is proportional to P s /Pr, where P s is the average power at the trans-
mitter as defined earlier. For P s proportional to b~ v , the function
P s /P' r has a maximum at b = B I - - 1 I (J 4 /J 3 ) 1/2 or b = 0.42B if v
Table II — Values of J n
Jl
0.5628
J,
0.0868
J,
0.0304
J,
0.0143
J,
0.0079
Jl
0.0048
HIGH-SPEED DIGITAL LIGHTWAVE COMMUNICATION 1369
= %. The latter condition results in a degradation of receiver sensitivity
(or an equalization penalty) of 2.4 dB over the values given in Fig. 1.
A faster led would reduce this penalty, but would provide less output
power and a smaller overall signal margin.
Chromatic and intermodal dispersion in the fiber also influence the
signal. Neglecting nonlinear, coherence and waveguide effects, we can
model the fiber transfer characteristic by the product of two filter
functions H c and Ha, one describing material dispersion and the other,
intermodal delay differences. For a given transmission distance z, the
material effect can be determined in advance for typical led systems
and can be compensated for in the receiver filter. At the wavelength of
minimum material dispersion (—1.3 jum), the transfer function attrib-
utable to material dispersion is 2
H c (f) = (1 + iirfz^/WcX 2 )- 1 ' 2 (5)
for a Gaussian led spectrum having an rms width a = 0.425 fwhm.
For typical InGaAsP leds with a doping density of 3 X 10 18 cm -3 , the
rms width is a — A 2 /40 /nm, where A is in /mi. 14
Figure 2 shows the increase in the required receiver power as a
result of material dispersion in fibers plotted against the product of bit
rate B and transmission distance z. The led bandwidth b is used as a
parameter. The penalties associated with transfer functions given by
eqs (1) and (5) add nearly linearly.
Present fiber fabrication processes are not sufficiently precise to
provide the desired control of intermodal delays (less than ~0.5 ns/km
total). Thus, unable to anticipate the resulting baseband characteristics
precisely, we must also allow for an equalization penalty due to
intersymbol interference in addition to that caused by first-order
equalization in the receiver. To estimate the latter effect, we use the
simple functional dependence
H d (f) = (\ + iSzf/Qy l (6)
following the usual notation of a bandwidth-distance product Q defined
at \H d \ - % (3 dB optical). The additional penalty factor [1 + 3(J 4 /
J3) (zB/Q) 2 ] 1/z can be calculated in a manner similar to that described
above for the source.
III. EXPERIMENTAL LEDs AND PIN PHOTODIODES
The high-radiance light-emitting-diode structure used in this work
is shown in Fig. 3. The diode was a small-area, etched-well surface
emitter, described previously, 15 made from a double-heterostructure
wafer of InP/Ini-^Ga^AsyPi-^/InP grown by liquid-phase epitaxy. The
composition of the quaternary active layer (x = 0.36, y = 0.80) was
chosen to produce diodes with an output wavelength of 1.3 jum ± 0.05
1 370 THE BELL SYSTEM TECHNICAL JOURNAL, OCTOBER 1 980
1000 2000
z-B IN KILOMETER x MEGABIT PER SECOND
3000
Fig. 2 — Increase in optical signal power required at the receiver as a result of material
dispersion in germanosilicate multimode fibers plotted vs product of distance z and bit
rate B; b is the led modulation bandwidth.
/urn, under room-temperature ambient conditions, at an operating
current of 100 mA dc. This current corresponds to a current density of
about 5 kA/cm 2 in the primary emitting area, which was about 50 /im
in diameter. The active quaternary layer, 1 to 1.5 /on thick, was doped
p-type with zinc to a relatively high carrier concentration of 2 to 4 X
10 18 /cm 3 . The 1.3-/im wavelength output into the air without antire-
flection coatings or lenses was about 1 mW at 100 mA. A typical
multimode fiber (50-/im core diameter, 0.23 n.a.), butt-coupled without
index matching, collected 10 /xW of that power. The spectral width
(fwhm) at 25°C and 100 mA was 1145A. The electrical modulation
bandwidth was ~170 MHz.
Temperature characteristics of the diodes were measured in the
range of 20°C to 100°C, and the output power was found to follow
approximately the relation
PT = P exp[-(T-r )/T c ], (7)
where P is the power output at T a (300°K) and the characteristic
temperature, T c , is 80° K. These results are plotted in Fig. 4, where it
HIGH-SPEED DIGITAL LIGHTWAVE COMMUNICATION 1371
CONTACTS
w/mr//.
1.3 -Aim LIGHT OUT
itALU*
- * W///////M/Mm%//,M.
. — InP
, - __- InGaAsP
■ InP
SiN
Fig. 3 — Schematic cross section of the 1.3-/un led.
can be seen that the output was reduced by 1.5 dB for a temperature
rise from room temperature to 55 °C. The thermal impedance was 80
to 100° C/W. The peak emission wavelength was shifted 380A at a rate
of ~5A/°C toward longer wavelengths by the same temperature in-
crease. The spectral width varied approximately as A 2 . The electrical
1.0
1-
=3 0.8
—
_
o
>
1- 0.6
"~~~>~»^^^ —
_j
w
"" 0.4
InGaAsP LED
-
0.2
-
1 1 1
1
20 40 60 80
TEMPERATURE IN DEGREES CELSIUS
100
Fig. 4 — Temperature dependence of wavelength, spectral width (fwhm) and output
of the 1.3-pm InGaAsP led.
1 372 THE BELL SYSTEM TECHNICAL JOURNAL, OCTOBER 1 980
modulation bandwidth, however, remained unchanged from 20 to
100°C. The practical operating life of these specific diodes is unknown,
but at this writing, similar unencapsulated diodes have been operating
at 100 mA dc in a room-temperature laboratory environment for over
23,000 hours with negligible degradation.
Photodetectors for the experiment were pin photodiodes made with
In^Gai-jAs lattice matched to InP (x = 0.53). They are described in
more detail elsewhere. 16 This ternary composition has a bandgap
energy of 0.75 eV, corresponding to a long-wavelength cutoff near 1.65
/mi. The attainable leakage current of p-n junctions in this material
has proved to be one to two orders of magnitude less (better) than
that associated with similar junctions in germanium, but otherwise the
photosensitive properties of the two materials are rather similar.
The InGaAs was grown by a conventional lpe technique to a
thickness of 6 to 8 /mi on a (100) n-type InP substrate. The epitaxial
layer was not intentionally doped and, with a suitable bake-out 17 of
the boat and starting materials before growth, the residual n-type
carrier concentration of the ternary layer was 1 to 2 X 10 15 /cm 3 . This
relatively low impurity concentration permits the fabrication of diodes
with low capacitance and depletion layer widths of 2 to 3 /xm at low
bias voltages. 1617
The photodiodes are illustrated schematically in Fig. 5. The p-n
junction was prepared by diffusion of zinc to a depth of 2.5 to 3.5 fan
into the grown InGaAs layer. After contacts were applied, small mesas
with a diameter of about 90 /un were formed by chemical etching and
windows about 250 jinn in diameter were opened in the substrate
CONTACTS
r~ ___
13=^
.InGaAs
EPOXY
1.3-/xm LIGHT IN
Fig. 5 — Schematic cross section of the back-illuminated long-wavelength pin photo-
diode.
HIGH-SPEED DIGITAL LIGHTWAVE COMMUNICATION 1373
metallization, also by chemical etching, to permit unobstructed illu-
mination of the junction through the substrate. This back-illuminated
configuration takes advantage of the transparency of the InP substrate
to minimize absorption loss and surface recombination and, at the
same time, to achieve minimum dark current and minimum parasitic
capacitance: there is no need to enlarge the junction area to provide
room for contacts or contacting pads. The diodes were operated
without an anti-reflection coating, and the external quantum efficiency
approached the transmission coefficient of 0.7 expected at the interface
between air and InP (refractive index 3.4).
The 90-/xm mesas had measured junction capacitances of 1 to 1.5 pF
at zero bias and 0.3 to 0.5 pF at 10 V reverse bias, where the depletion
width was about 3 /im and the peak electric field was estimated to be
6.7 X 10 4 V/cm. The best units had dark currents between 1 and 2 nA
at 25 °C and 10 V reverse bias; the variation of the dark current with
temperature was exp(-E g /2kT) up to 100°C (see Fig. 6). The dark
current of the units used in the present repeater experiment ranged
io-
i<r° -
< 10"
icr s -
10^
1 1
InGaAsPHOTODIODE
NO. 139
DIAMETER 100/im ^^^
1
-
140 °C^-
*"■"*
„*■"*" 100 a c^*-^^
< ^-"" 80 'C^ — -"■""^
^^■^^"^ 60 °C^-
^■"^^ 40 "C^^^^ ^.
^ ^"^ 25 °C^ — "^"^
1 1
1
5 10 15
REVERSE VOLTAGE (V)
20
Fig. 6 — Dark-current characteristics of one of the best InGaAs diodes in the temper-
ature range between 25 and 100°C.
1374 THE BELL SYSTEM TECHNICAL JOURNAL, OCTOBER 1980
RECEIVER
TRANSMITTER
<j
i)
PHASE-
LOCKED
LOOP
P.N^
AMPLIFIER
! AMPLIFIER
'/
EQUALIZER
/
FILTER
DECIDER
~\~
tf
i
AMPLIFIER
■>
AGC
Fig. 7 — Block diagram of the optical repeater.
from 5 to 100 nA at room temperature. The reverse breakdown was 75
to 100 V at 10-juA bias current. The parasitic capacitance of the
modified silicone-potted "pill" package used in these experiments was
0.1 to 0.15 pF. The measured rise times of the better units were less
than 0.2 ns.
IV. REGENERATOR DESIGN
To test the feasibility of the system configuration described earlier,
we assembled two experimental regenerators designed to operate at
44.7 and 274 Mb/s. Figure 7 shows a block diagram of the essential
functions. The transmitter module was the same for both regenerators
and consisted of a two-stage GaAs fet power amplifier driving the led
previously described. The amplifier produced pulses of 200-mA peak
current and 500-ps rise time. With this design, it was possible to
generate pseudo-random optical pulse patterns at bit rates up to 365
Mb/s. Figure 8 shows the transmitter output at 274 Mb/s. The average
power launched into a fiber of 50-/xm core diameter and 0.23 n.a. was
about 5 jtW.
The receiver module consisted of the linear amplifier and filter,
decision, and timing recovery circuits. Except for the timing recovery
function, the 274-Mb/s design made use of high-speed integrated
circuits based on the microwave isolated-junction technology. The
timing recovery circuit used was a standard part of the Western
Electric regenerator designed for the T4M coaxial transmission sys-
tem. 18 The linear channel consisted of the optical detector, a GaAs fet
preamplifier described below, an equalizer, a 60-dB booster amplifier
with automatic gain control, and a multistage lc filter designed to
produce the desired noise characteristic. 11 The booster and regenerator
portions of the 44.7-Mb/s regenerator were standard parts presently
used in the FT3 lightwave transmission system. 19
The front-end portions of both regenerators consisted of the InGaAs
HIGH-SPEED DIGITAL LIGHTWAVE COMMUNICATION 1375
detector previously described, a low-noise dc-coupled feedback ampli-
fier as shown in Fig. 9 and an rc equalizing circuit. The three-stage
amplifier was comprised of a GaAs fet and two complementary
microwave transistors. 11 The critical circuit parameter was the input
capacitance Ct, which consisted of the photodiode capacitance (0.5
pF), the fet gate-source capacitance (0.5 pF), and the parasitic circuit
capacitance (0.5 pF). With Ct = 1.5 pF and g m = 30 mS, the perform-
ance resulting from fet channel noise can be found in Fig. 1.
The extraordinary speed of the microwave transistors used {f c = 4
GHz), as well as a special computer-aided circuit design process, made
it possible to construct a feedback amplifier useful at bit rates up to
several hundred megabits per second. This approach assured wide
amplifier bandwidth, wide dynamic range, and a single-pole character-
istic which minimized the subsequent equalization required. Figure 10
shows the test package with coaxial connections. The open-loop gain
A and the feedback resistance Rl were chosen as indicated in Table
III. The objective was to maximize the bandwidth of the feedback
amplifier, which is roughly proportional to A/R l Ct, without adding
significant front-end noise. The first term in eq. (2) describes the noise
Fig. 8 — Eye diagram at the transmitter output (B = 21 A Mb/s).
1376 THE BELL SYSTEM TECHNICAL JOURNAL, OCTOBER 1980
Fig. 9 — GhAsfet preamplifier circuit diagram.
contribution due to R L ; for the two cases listed in Table III, the noise
contributions are shown in Fig. 1 by the two crosses.
The gate leakage current of the GaAs fets was a function of
temperature and gate bias voltage. For an operating bias V G s of —0.5
V, the best units had a leakage current of 1 to 2 nA which increased to
5 to 10 nA at 70° C. 20 As in the case of the detector diode, there was a
significant spread in performance, but most of the results described
below were obtained with units not specially selected. At bit rates
below 100 Mb/s, the 1//" noise component of the GaAs fet can impair
the sensitivity of the pin-fet receiver. We found this degradation to
be negligible to bit rates as low as 30 Mb/s in all units employed in our
receiver studies.
V. EXPERIMENTAL RESULTS AND CONCLUSIONS
Two system experiments were performed, one at each bit rate, by
connecting the transmitter and receiver modules shown in Fig. 7 via
multimode test fibers. Pseudo-random bit patterns were transmitted
at the design rate and the system was tested for bit errors. Figure 11
shows the results. The solid lines are best fits to the data points.
For comparison with theory, one can compute P r from eqs. (2) and
(3) using g m = 50 mS, C T = 1.5 pF, R L from Table III and I L = 75 nA
as measured for the combined leakage current of the fet and the
photodiode. The external quantum efficiency of the detector in the
44.7-Mb/s receiver was 0.50, while for the 274-Mb/s receiver tj was
0.66. With these values, we calculate an expected receiver sensitivity
of -49.7 dBm at 44.7 Mb/s and -39.8 dBm at 274 Mb/s, on the
HIGH-SPEED DIGITAL LIGHTWAVE COMMUNICATION 1377
Fig. 10— Front-end module with coaxial test connections.
assumption that the led and the fiber have infinite bandwidths.
Actually, the led bandwidth was 170 MHz, while the fibers used had
bandwidths (3-dB optical) of 45 MHz in the 44.7-Mb/s experiment and
200 MHz in the 274-Mb/s experiment. As explained in Section III,
these conditions lead to a modification of the J- values in eq. (2). If one
uses eqs. (1), (5), and (6) to compute the penalty factors, the expected
sensitivities become -48.2 dBm at 44.7 Mb/s and -36.8 dBm at 274
Mb/s.
The discrepancies from the measured data shown in Fig. 11 are 1.5
and 0.8 dB. We ascribe the difference at 44.7 Mb/s to induced gate
noise and to the uncertainty in determining the actual amount of
leakage current, which in our experiments fluctuated with humidity
and near-term operating history. The discrepancy at 274 Mb/s is
probably a consequence of intersymbol interference which was ne-
glected in our computation. However, the relatively good agreement
1378 THE BELL SYSTEM TECHNICAL JOURNAL, OCTOBER 1980
10"
i= 10-
10"
io-
,_^_--46.7dBm
J I I I I L
J L
-50
-45 -40
AVERAGE OPTICAL POWER IN dBm
-35
Fig. 11 — Error rate vs average optical input power measured at 44.7 and 274 Mb/s.
gives us confidence that the equalization in the receiver can compen-
sate for the known bandwidth limitations of the optical channel.
The remaining question of interest concerns the expected repeater
spacing possible with led-pin systems operating at speeds above 40
Mb/s. Based on current research, the answer will naturally be con-
servative, but nevertheless it will provide an interesting benchmark.
Allowing for component variations and elevated temperatures, we
assume g m = 30 mS, Ct = 1.5 pF, and h = 200 nA, and compute the
average optical power P r from eqs. (2) and (3), taking into account the
led modulation characteristics and fiber dispersion due to chromatic
and intermodal delays; e.g., we use the appropriate J-coefficients in
accordance with the actual transfer conditions as stipulated by eqs.
Table III — Pre-amplifier characteristics
Open Feedback Band-
Bit Rate Loop Resistance width
Mb/s Gain kSl (MHz)
44.7
274
30
12
500
30
3
50
HIGH-SPEED DIGITAL LIGHTWAVE COMMUNICATION 1379
(1), (5), and (6). This computation converts bandwidth limitations into
an equalization penalty at the receiver. Lacking more specific infor-
mation on intersymbol interference, we allow for an equalization
penalty in decibels equal to the ratio zB/Q, where Q is the bandwidth-
distance product resulting from intermodal delays. (This assumption
is based on limited experimental experience and deserves further
study.) In addition to these receiver sensitivity penalties, there are
fixed margin (power-loss) allocations at the transmitter and the re-
ceiver as listed in Table IV.
The average led output into a typical fiber (50-/im core, 0.23 n.a.)
is assumed to be —13 dBm between and 50 MHz, and to decrease by
2 dB between 50 and 100 MHz and by 3 dB/octave thereafter. The
led bandwidth is chosen for maximum signal margin P a /P r . This ratio,
reduced by penalties and allocations, determines the transmission
distance possible with a cable having a given over-all loss. Since P s /P r
is a function of z because of eqs. (5) and (6), we must seek an iterative
solution to determine z.
Some representative results are shown in Fig. 12. Cable loss and the
bandwidth- distance product associated with intermodal dispersion are
the two parameters used. We make the conservative assumption that
the bandwidth decreases linearly with length. The transmission dis-
tances achievable range from 24 km at 44.7 Mb/s to 8 km at 274 Mb/
s for cable having a loss of 1 dB/km and a bandwidth-distance product
in the 1000-MHz • km range. The good agreement with experimental
results seems to justify the underlying design rules. The influence of
induced gate noise and the determination of residual intersymbol
interference need further study.
Three interesting conclusions can be drawn from the calculation
that resulted in Fig. 12:
(i) Loss limits the achievable repeater spacing more critically than
bandwidth. For a given application, the bandwidth specification and
the associated equalization penalty allocation are design variables to
be determined ultimately by economic considerations.
(ii) The "loss-limited" repeater spacing of led-pin systems varies
Table IV — Fixed margin allocations
Transmitter:
30 °C temperature rise
1.5 dB
Output decrease to end of life
1.5 dB
Connectors
1 dB
Receiver
Feedback resistance
1 dB
Connectors
1 dB
Margin
3 dB
Total
9 dB
1380 THE BELL SYSTEM TECHNICAL JOURNAL, OCTOBER 1980
30
20
<
uj
°r. 10
III I
N. N. 1dB/km
— ^v ^v
^\2dB/km y\
^ S ^^>». 2000 M Hz- km r'' N^ N.
1000MHz-km<^
44.7 Mb/s
274 Mb/s
Vlli I
V
40
60
80 100 200
BIT RATE IN MEGABITS PER SECOND
400
Fig. 12 — Expected repeater distance versus bit rate for led-pin repeater at 1.3 /un;
installed cable loss and bandwidth -distance product of fiber are the parameters.
approximately as the inverse of the bit rate in the range between 100
and 300 Mb/s. Therefore, since this implies a constant bit-rate-distance
product, the bandwidth requirements are nearly the same for all these
bit rates.
(Hi) For a fiber bandwidth between 1000 and 2000 MHz -km and a
spliced cable loss of — 1 dB/km and allowing a receiver equalization
penalty of 2 to 4 dB, it is possible to build high-speed (>100 Mb/s)
led-pin systems with repeater spacings given in Fig. 12. Since the
equalization penalty increases very rapidly with repeater spacing, an
increase in source power (e.g., from the use of a laser) would permit
larger margin allocations and penalties but would not lead to substan-
tially longer repeater spacings in high-speed multimode systems.
VI. ACKNOWLEDGMENTS
We thank W. R. Northover, M. Saifi, and J. W. Shiever for the test
fibers used, W. O. W. Schlosser and S. H. Wemple for their help with
the GaAs fets, W. Kruppa for providing the microwave transistors
HIGH-SPEED DIGITAL LIGHTWAVE COMMUNICATION 1381
and integrated circuits, and S. E. Miller for guidance and encourage-
ment through all phases of this work.
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1382 THE BELL SYSTEM TECHNICAL JOURNAL, OCTOBER 1980